The present invention relates generally to amplifiers, and more particularly to improved over-voltage protection for differential amplifiers.
Most high voltage operational amplifiers use some type of input over-voltage protection, only a few of which are “robust”, for example those using a lateral-PNP input stage or simple high voltage JFET input stage. A typical bipolar differential input stage as shown in FIG. 1, for example, is protected against differential input voltages of up to about 0.7 volts by means of a pair of diodes D1 and D2 coupled in opposite directions across the input limiting differential voltage, but this protection is achieved at the cost of a very large input current flowing through the protection diodes D1 and D2 if the input differential voltage magnitude exceeds about 0.7 volts.
More complicated input stages like the common-drain-common-base topologies shown in FIGS. 2 and 3 usually do not have input protection diodes as shown in FIG. 1, but they have an internal diode clamp protecting the bipolar portion of the input stage. This type of input stage relies on the high breakdown voltage of JFETs and provides good DC performance (e.g., low input bias current) even for a differential input voltage value approaching the full supply voltage. However, a transient response problem exists in operational amplifiers of the type using common-drain-common-base input stages which utilize an over-voltage protection clamp circuit. The transient response problem is especially pronounced when large input transistors with large parasitic capacitances are utilized.
The input stage circuits 1A and 1B shown in FIGS. 2 and 3, respectively, use common-drain-common-base topology and require protection from a large differential voltage between the emitter of transistor Q3 and the emitter of transistor Q4. FIG. 2 shows a complete operational amplifier including a conventional input stage 1A and a conventional output stage 2. Conventional output stage 2 can be used in conjunction with various improved input stages (subsequently described) of the present invention. During certain modes of operation of the operational amplifier, for example during slewing operation, large input differential voltages may appear. If a positive, high slew rate input signal Vin+ is applied to the gate of transistor J1 (i.e., to the non-inverting input of the operational amplifier) in FIG. 3, the emitter voltage of transistor Q3 also will rise as fast as the input signal Vin+. However, the gate voltage Vin- of transistor J2 would be coupled by a feedback element to the output of the operational amplifier, so the emitter voltage of transistor Q4 would only rise as fast as the slew rate of the output of the operational amplifier. If the slew rate of the operational amplifier output stage is much slower than the input signal slew rate of Vin+, there will be a large voltage difference between the emitters of transistors Q3 and Q4. In this example, the base-emitter junction of transistor Q4 will be highly reverse biased. This is problematic because typically the base-emitter junction of an integrated circuit bipolar transistor can not be reverse biased by more than about 2 to 3 volts without permanently damaging the transistor.
A typical clamp circuit that can solve the foregoing problem of permanently damaging transistors Q3 and Q4 includes a pair of strings of diodes D11 . . . D1n and D21 . . . D2n coupled between the emitters of transistors Q3 and Q4, as shown in FIG. 3. This clamp circuit limits the voltage difference between the emitters of transistors Q3 and Q4 so as to limit the amount of reverse bias voltage across the base-emitter junction of transistor Q4 for positive slewing (or Q3 for negative slewing) to a value below its maximum allowed value. (Note that this type of clamp circuit cannot be used if bipolar transistors are used as the input transistor pair, because it may excessively reverse bias the base-emitter junction of the bipolar transistor corresponding to JFET J2. In contrast, if JFETs are used as the input transistors, their gate-source junctions typically can withstand the amount of reverse bias that the clamp circuit causes.) Unfortunately, the clamp circuit of FIG. 3 has the problem that it causes charging of the large gate-source parasitic capacitors Cp1 and Cp2, which can degrade the output slew rate of the output stage of the operational amplifier. During slewing in the positive direction, a large differential input signal Vin=Vin+-Vin− is applied to input stage 1B, causing the diode clamp circuit D11 . . . D1n to turn on and limit the amount of reverse bias voltage across the base-emitter junction of transistor Q4. At the same time, the gate-source junction of input transistor J2 is reverse biased and the parasitic capacitor Cp2 is charged up to the value of the input signal Vin=Vin+-Vin− minus the voltage drop across the clamp circuit. As the slew-limited inverting input voltage Vin− is slowly slewing up, the source voltage of input transistor J2 tracks it, thereby increasing the emitter voltage of transistor Q4 and turning it on. At that moment the charged-up parasitic capacitor Cp2 begins to discharge into the emitter of transistor Q4. If parasitic capacitor Cp2 is large, the resulting parasitic capacitance discharge current (ICpar) discharged through transistor Q4 also is large. At this point, it should be noted that in the ideal case, in which the parasitic capacitance is small, transistors J1, Q3, Q5 and Q6 are conducting maximum current during positive slewing to produce the recharging current Iout1, whereas transistors J2 and Q4 should be completely off. But due to the above described parasitic capacitor Cp2 discharge, transistor Q4 actually steals a substantial fraction of the current Iout1, which reduces the slew rate of the amplifier.
In the case of large gate-source parasitic capacitance Cp2 as shown in FIG. 3, the associated parasitic charging current ICpar described above can approach the magnitude of the amount of tail current I1*A1 available to the differential input transistors J1 and J2. (Here, A1 is the current gain of transistors Q3 and Q4 (beta). In practice, for better amplifier performance the current gain A1 is limited to a lower value by means of additional scaling diodes such as diode-connected transistors Q3B and Q4B in subsequently described FIGS. 5 and 6. In that case the current gain A1 is equal to the ratio of the emitter area of transistor Q4 (Q3) to the emitter area of transistor Q4B (Q33B), which is better controlled than the transistor current gain beta.) In the case in which ICpar is large, Iout1, which is the difference in the current through transistors Q6 and Q4 (I1*A1-ICpar), becomes substantially smaller than I1*A1. As a result, the compensation capacitor Ccomp (see FIG. 2) of the operational amplifier output stage 2 is charged at a lower rate than in the ideal case. This degrades the slew rate of the operational amplifier in FIG. 2, the slew rate being S=(I1*A1-ICpar)/Ccomp, especially when the input stage receives a large input swing that causes the above-mentioned clamp circuit to turn on and begin the input parasitic capacitor charge-discharge process.
Another problem associated with the above described parasitic capacitance discharging current is high differential input capacitance and input error caused by the resulting high parasitic capacitance recharging currents reacting with the input signal source impedances. This problem has two aspects. The first aspect is just the amount of charge flowing through the input during an input voltage transient, wherein the larger the voltage change across the parasitic capacitance Cp2, the greater the amount of parasitic capacitor discharging current. The second aspect is the nonlinearity of the phenomenon. On the positive input signal edge, the source voltage of the input JFET J1 follows its gate voltage, and the VGS modulation and Cp1 recharge current are low, whereas the Cp2 recharge current is large. On the negative input signal edge the resulting large amplitude source voltage of input transistor J1 does not follow its gate voltage and is determined by the other input voltage minus the voltage drop across the diode clamp circuit. That causes a substantial recharge current flowing through the gate of the input JFET, i.e., the input of the amplifier. So the input current and the product of its reaction with the input signal source impedance is substantially different for positive and negative edges of the input signal, and this is a nonlinear “external” effect of amplifier which adds to the internal nonlinearity of the amplifier. Another possible problem is that if on a positive edge of the input signal the Cp2 recharge current is very large, it can exceed the IDSS specification value of input transistor J1 and its gate-source p-n junction will be forward biased, which can cause very large settling times, large transient bias currents, etc.
Another problem of over-voltage protection of an input stage with the diode clamp circuit D11 . . . D1n, D21 . . . D2n of FIG. 3 is that the maximum input voltage is limited by the VGS breakdown voltage of the input JFETs. This may be of concern because wafer fabrication techniques which improve JFET performance often cause reduction of the VGS breakdown voltages of JFETs. Consequently,. with such reduced breakdown voltage it becomes impossible to provide high (i.e., full supply voltage) absolute maximum differential voltage specifications when using diode clamp protection circuits having low clamp voltage.
There is an unmet need for an input stage which avoids large modulation of voltage across the parasitic capacitors of the input transistors (which large modulation generates large parasitic currents that degrade performance of the amplifier) while also effectively preventing damage caused by excessive reverse bias voltage across emitter-base junctions of transistors in the input stage.
There is an unmet need for an input stage which avoids degrading transient response of the amplifier and, in particular, the slew rate of an amplifier caused by unwanted currents through parasitic capacitors associated with input transistors.
There also is an unmet need for an input stage with reduced input errors caused by charging currents through parasitic capacitances of input transistors of the input stage.
There is an unmet need for an input stage which allows for using high-performance transistors but with lower breakdown voltage by avoiding large modulation of gate-to-source voltage of the input transistors while still providing high (full supply voltage) absolute maximum values of the input differential voltage.
There is an unmet need for a bipolar input stage which maintains low input bias current with substantially higher than 0.7 volt maximum input differential voltage (which ideally is equal to the full supply voltage) wherein at the same time the base-emitter junctions of the input transistors are fully protected from high input differential voltages.